An NMR system generally includes transceiver circuits for transmitting signals to a tested sample and receiving echo signals therefrom and a processor for analyzing the echo signals in order to obtain imaging and/or material information of the sample. Recently, significant efforts have been devoted to miniaturize traditional NMR systems, in particular NMR transceivers. The numerous advantages of miniaturization include low cost, portability, and the fact that a micro-coil tightly surrounding a small size sample increases the signal quality.
The practical design and construction of miniaturized NMR transceiver circuits, however, may present a number of difficulties. For example, designing an integrated mixer with sufficient power and area efficiency that also meets multiple design requirements (such as linearity, bandwidth, dynamic range, noise, gain mismatch, and offset) can be difficult. This is especially true at the lower supply voltages seen in modern semiconductor devices. If these often-conflicting design requirements are not met, the overall performance of the receiver will suffer.
For example, FIG. 1A illustrates a conventional quadrature receiver 100 including I/Q mixers 102 for down-converting the input signal to a lower frequency and generating both in-phase (I) and quadrature (Q) output signals. Down-converting the frequency of the input signal helps relax the requirements of the analog-to-digital converter (ADC) used to digitize the signal. Integrated mixers typically fall into two categories, depending on whether the mixer core circuitry is active or passive.
An exemplary active mixer is the conventional Gilbert Cell mixer 112 shown in FIG. 1B. In the Gilbert Cell mixer 112, a radio-frequency (RF) input signal is first converted to a current using the Q1 devices (e.g., transistors) 114 and subsequently mixed with a local oscillator (LO) signal through the Q2 devices 116. If inputs to the RF and LO ports include discrete sine waves, an intermediate frequency (IF) output across the load resistors 118 may include components at frequencies of (fRF−fLO) and (fRF+fLO). The up-converted signal at the frequency of (fRF+fLO) may be filtered using a baseband filter following the mixer 112; this leaves a desired baseband output at the frequency of (fRF−fLO). One issue with the Gilbert Cell mixer 112 is the limited headroom and poor linearity resulting from the stacked devices Q1, Q2, and the tail current source 120. In addition, the active mixer 112 may generate 1/f noise, which degrades the signal-to-noise ratio of the input signal when it is converted to a baseband frequency. These issues are particularly concerning in modern complementary metal-oxide-semiconductor (CMOS) processes, which have lower supply voltages and larger 1/f noise compared to bipolar devices.
On the other hand, passive mixer architectures have recently become more popular as they offer improved linearity and decreased 1/f noise compared to their active counterparts, especially in lower-voltage CMOS processes. FIG. 1C depicts a typical exemplary passive mixer architecture 132; this mixer is classified as a current-mode mixer since the transconductance stage 134 at the input converts the RF input signal into a current before it is mixed with the LO signal using mixer switches S1 136 and S2 138. The LO signal in the passive mixer 132 is a rail-to-rail square wave which turns one set of the switches 136, 138 fully on and the other set fully off based on the polarity of the signal. This feature differentiates the passive mixer 132 from the active mixer 112 that has the mixer switches biased in the active region. The output current of the passive mixer switches 136, 138 flows through feedback resistors 140 of the output transimpedance stage 142 to provide a voltage. Capacitors may be placed in parallel with the feedback resistors 140 to filter out the undesired up-converted signals.
Because stacked devices are not needed in the transconductance stage 134 (and the voltage change at the output of the transconductance stage 134 is small in the passive-current mode mixer 132), the linearity of the passive-current mode mixer 132 is generally improved compared to the Gilbert Cell mixer 112. In addition, the 1/f noise is improved in the passive-current mode mixer 132 because no direct current flows through the switches 136, 138.
The current-mode mixer architecture 132, however, cannot easily reject direct-current (DC) offsets from the transconductance stage 134 because the signal is current-mode. As a result, conventional passive current-mode mixers require complex trimming routines to inject current into one of the switching nodes that operate at RF frequencies. Further, because the gain of the current-mode mixer 132 depends on the voltage-dependent transconductance stage 134, non-linearities may still occur. In addition, it may be difficult to match mixer gains at RF frequencies without loading the preceding stage. The latter issue is a particular concern in the quadrature receiver architecture shown in FIG. 1B.
Accordingly, there is a need for an approach that allows mixers in a receiver to down-convert the received signals to baseband frequencies with improved linearity and decreased 1/f noise while avoiding mismatches in gain, bandwidth, and offset between the mixers.